The present invention is related to operational amplifier design. More specifically, the present invention teaches a variety of linear and multi-sin h transconductance circuits each well suited for use as a low-distortion input stage in an operational amplifier.
In the design of operational amplifiers, it is important to provide a highly linear (i.e., low distortion), low noise amplifier capable of wide bandwidth operation. Bandwidth limitations, noise, and distortion can arise at any stage within the operational amplifier, but for present purposes the focus is upon the input stage. The typical input stage is a transconductor or transconductance circuit operable to convert an input voltage signal into an internal current signal more suitable for amplification by the output stage. Hence, the defining feature of the transconductance circuit is its voltage to current transfer function.
Prior Art FIG. 1 illustrates the prototypical input stage transconductor 10, i.e., a differential transistor pair. The transconductor 10 includes a pair of transistors Q1 and Q2 whose emitters are coupled to a bias current source IDC that provides xe2x80x9ctailxe2x80x9d current for the transconductor 10. The differential voltage input pair VIN+ and VINxe2x88x92 drive the bases of the transistors Q1 and Q2, essentially steering the resulting differential current pair IOUT+ and IOUTxe2x88x92 to a common ground reference 20. As will be appreciated, the voltage to current transfer function of the differential pair transconductor 10 is ideally a hyperbolic tangent (tan h) function.
While widely applicable and well suited for certain applications, the transconductor 10 suffers many shortcomings. When used within an amplifier having a capacitive feedback loop, as is often the case, the transconductor 10 is extremely limiting on the slew rate. (An amplifier""s slew rate defines the maximum rate of change in voltage across the input and output terminals of the amplifier.) Specifically, the total current available to charge the feedback loop compensation capacitor CC is limited by the so-call xe2x80x9ctail currentxe2x80x9d of the differential pair, i.e., the bias current IDC.
For the present analysis, it is fair to assume that the slew rate is equal to IDC/CC. Hence to improve the slew rate, one must decrease CC and/or increase IDC, both of which are undesirable for a variety of well known reasons. Additionally, the tan h transfer function of the differential pair transconductor 10 means that transconductor 10 is a non-linear, distortive circuit.
One common approach for addressing the slew rate limitations of the differential pair transconductor 10 of FIG. 1 is to use a class AB transconductance amplifier. Prior Art FIG. 2 illustrates one typical class AB amplifier 100 formed from a pair of differentially coupled diamond followers whose output emitters are coupled through a common load resistance RDGEN. Each diamond follower includes a pair of bias current sources IDC, and four transistors (one follower is made of transistors Q1-Q4, the other follower is made of transistors Q5-Q8).
The voltage to current transfer function of the class AB amplifier 100 without a common load resistance RDGEN (i.e., RDGEN=0) is ideally a hyperbolic sine (sin h) function. Prior Art FIG. 4 illustrates such an ideal transconductance of the class AB amplifier 100 (i.e., dIout/dVout) as a function of input voltage. As seen in FIG. 4, the ideal transconductance of the class AB amplifier 100 is non-linear at voltages close to zero, but fairly linear elsewhere. The transfer function of the class AB amplifier will vary for different values of RDGEN, but the non-linear characteristics are similar and related to the sin h function represented in FIG. 4.
In practice, the transconductance gain of the class AB amplifier 100 is set by the available bias current, the common load resistor RDGEN, and the nonlinear transconductance characteristics of the individual transistors. However, when RDGEN is large it dominates the nonlinear effects of the individual transistors, thereby improving the distortion characteristics of the class AB amplifier 100. Unfortunately, increasing RDGEN increases noise in the class AB amplifier 100 due to thermal noise of the resistor.
As mentioned above with reference to the differential pair transconductor 10 of FIG. 1, much of the non-linearity of transconductor 10 is due to the tan h nature of its transfer function. One well-known technique for linearizing differential pair transconductors is the so-called xe2x80x9cmulti-tan h technique.xe2x80x9d As will be appreciated, the key to the multi-tan h technique is the placement of multiple nonlinear tan h transconductors (i.e., differential pairs) along the input-voltage axis to achieve in combination a more linear transfer function.
Prior Art FIG. 3 illustrates a multi-tan h doublet 200 formed from two differential pairs Q1-Q2 and Q3-Q4 and two bias current sources IDC. Positive and negative offsets are introduced by forming each differential transistor pair with a gain imbalance. Specifically, a positive offset is introduced into the differential pair Q1-Q2 by forming transistor Q1 with a gain A that is greater than unity, and transistor Q2 with a gain of substantially unity. Likewise, a negative offset is introduced into the differential pair Q3-Q4 by forming transistor Q4 with a gain A that is greater than unity and transistor Q3 with a gain of substantially unity. Prior Art FIG. 5 illustrates the combined transconductance gain.
The multi-tan h transconductors do improve the distortion characteristics of an input stage, however the multi-tan h technique does not address the slew rate and other problems of the differential pair transconductor. Likewise, the class AB amplifier provides an improved slew rate, yet suffers from the nonlinearity about zero due to its sin h transfer function. What are needed are a variety of transconductance circuits that are highly linear with low noise, and having bandwidth characteristics not limited by slew rate.
The present invention teaches a variety of transconductance circuits such as transconductance circuits formed having a plurality of class AB transconductor amplifiers coupled in parallel. The class AB transconductor amplifiers have non-linear voltage to current transfer functions. Each class AB transconductor amplifier is designed with an offset chosen such that the individual nonlinear transfer functions are arranged along the input voltage axis to achieve a more linear transfer function for the combined transconductance circuit.
For example, a first embodiment of the present invention discloses a transconductance circuit characterized by a voltage to current transfer function, the transconductance circuit including a pair of class AB transconductance amplifiers coupled in parallel across differential input and output pairs. The first class AB transconductance amplifier has a positive offset. The second class AB transconductance amplifier has a negative offset. These negative and positive offsets are selected to improve linearity of the voltage to current transfer function of the transconductance circuit. It is contemplated that offsets may be of equal or differing magnitudes.
In certain embodiments, the class AB transconductance amplifiers are formed from a pair of differentially coupled diamond followers and a common load resistance RDGEN. Each diamond follower has four transistors and two bias current sources. The transconductance of each class AB amplifier is thus a function of transistor gain, the available bias current and the common load resistance RDGEN.
The present invention further teaches an operational amplifier having an input stage and a second stage (e.g., gain stage or output stage) coupled in series. The input stage, characterized by a voltage to current transfer function, includes a plurality of class AB transconductance amplifiers coupled in parallel. Each of the class AB transconductance amplifiers has an offset selected such that the combination tends to improve the linearity of the transfer function of the input stage.
Many embodiments of the present invention will involve coupling directly the outputs of parallel class AB transconductance amplifiers. However, such a step is not necessary to accomplish the improvements in linearity contemplated by the present invention. For example, the parallel class AB transconductance amplifiers often represent one stage in a circuit having multiple stages serial and/or parallel coupled. So, the output of a first class AB transconductance amplifier may drive one subsequent stage, while the output of a second class AB transconductance amplifier may drive another subsequent stage. These two subsequent stages are later combined (either directly or indirectly) thereby providing increased linearity at a subsequent stage (such as the output) of the multiple stage circuit.
As implied in the preceding paragraph, other circuitry may be coupled in parallel with the class AB transconductance amplifiers. This results in a transconductance stage providing the improved linearity of parallel coupled, properly offset class AB transconductance amplifiers, together with the electrical characteristics introduced by the additional parallel circuitry.